Active Load

Active (dynamic) load is a component or a circuit behaving as a current-stable nonlinear resistor. This term may refer to a component of circuit design, or to a type of test equipment.

Circuit Design

In circuit design, an active load is a circuit component made up of active devices, such as transistors, intended to present a high small-signal impedance yet not requiring a large DC voltage drop, as would occur if a large resistor were used instead. Such large AC load impedances may be desirable, for example, to increase the AC gain of some types of amplifier. Most commonly the active load is the output part of a current mirror and is represented in an idealized manner as a current source. Usually, it is only a constant-current resistor that is a part of the whole current source including a constant voltage source as well (the power supply V_CC on the figures below).

Common Base Example

In Figure 1 the load is a resistor, and the current through the resistor is determined by Ohm's law as:

.

As a consequence of this relation, the voltage drop across the resistor is tied to the current at the Q-point. If the bias current is fixed for some performance reason, any increase in load resistance automatically leads to a lower voltage for V_out. which in turn lowers the voltage drop V_CB between collector and base, limiting the signal swing at the amplifier output. (If the output swing is larger than V_CB, the transistor is driven out of active mode during part of the signal cycle).

In contrast, using the active load of Figure 2, the AC impedance of the ideal current source is infinite regardless of the voltage drop V_CC - V_out, which allows even a large value of V_CB. and consequently a large output signal swing.



Differential Amplifiers

Active loads are frequently used in op-amp differential input stages, in order to enormously increase the gain.

Practical Limitations

In practice the ideal current source is replaced by a current mirror, which is less ideal in two ways. First, its AC resistance is large, but not infinite. Second, the mirror requires a small voltage drop to maintain operation (to keep the output transistors of the mirror in active mode). As a result, the current mirror does limit the allowable output voltage swing, but this limitation is much less than for a resistor, and also does not depend upon the choice of bias current, leaving more flexibility than a resistor in designing the circuit.

Test Equipment

In the area of electronic test equipment, an active load is used for automatic testing of power supplies and other sources of electrical power to ensure that their output voltage and current are within their specifications over a range of load conditions, from no load to maximum load.

One approach to test loads uses a set of resistors of different values, and manual intervention. In contrast, an active load presents to the source a resistance value varied by electronic control, either by an analogue adjusting device such as a multi-turn potentiometer or, in automated test setups, by a digital computer. The load resistance can often be varied rapidly in order to test the power supply's transient response.

Just like a resistor, an active load converts the power supply's electrical energy to heat. The heat-dissipating devices (usually transistors) in an active load therefore have to be designed to withstand the resulting temperature rise, and are usually cooled by means of heatsinks.

For added convenience, active loads often include circuitry to measure the current and voltage delivered to the inputs, and may display these measurements on numeric readouts.

Fuente: http://en.wikipedia.org/wiki/Active_load

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

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Design and Analysis Techniques for Dynamic Current Mirrors

Today's world of electronics becomes more and more digital and therefore CMOS becomes the dominant technology. A CMOS process compared to a bipolar process offers several advantages, mainly a low power consumption which is important for portable systems powered by batteries or for large systems. Another point is the smaller device geometries, which increase the number of gates that can be packed on a single chip. Parallel to the increasing digital domain the analog field persists, because the origin of physical phenomenons and their perception is analog. Electronic sensors perceive these analog events and deliver a corresponding output signal. If this analog signal is not suitable for digital processing some analog preconditioning must be done [YEN82]. Combining both types of signal processing on the same chip to obtain the best possible performances is therefore desirable [MID84]. CMOS technology has led to a wide use of voltages and charge transfers for signal processing. As the feature sizes shrink a lower supply voltage is imposed, which reduces the dynamic voltage range of the circuits. Because process parameters are chosen to optimize digital performances, the analog functional blocks have to adapt themselves to the restricted voltage range. Current mode circuits offer a solution to these problems as they require only a baseline digital process and avoid many of the anticipated low voltage problems by operating in the current domain [HUG89]. Due to the non-linear current-voltage relationship the dynamic range of current-mode signals is larger than that of voltage-mode signals. Therefore the current mode approach can provide attractive and elegant solutions for many circuit and system problems [TOU90]. A ubiquitous elementary building block in most analog integrated circuits is the current mirror which is able to multiply and duplicate an imposed input current, that contains the information (bias or signal). The reproduced output current of is then available for any subsequent processing. Unfortunately, due to random process variations, transistor parameters are affected by a certain variation of the transconductance parameter and of the threshold voltage [SHY84], [LAK86]. Hence, the output currents of transistors which have been designed identically are different. These random variations, the so-called devices mismatch, are a major limitation for most accurate and precise current mode circuit applications. Another main limitation of CMOS circuits is the 1/f flicker noise of the MOS transistors. The standard technique to reduce this 1/f noise and the error due to mismatch is to increase the transistor area and to overwhelm the threshold mismatch with a high gate voltage overhead, which simultaneously increases the saturation voltage of the devices. The performance of a mirror with low saturation voltage can be improved by using lateral bipolar transistors [VIT83]. The resulting current error can be lower than 1%, but the major handicap is that only one type of mirror can be built (source or sink depending on the technology used). Special circuit techniques allow us to reduce the inherent noise and offset in MOS amplifiers, like the chopper technique [HSI81], [ENZ89], and the auto-zero technique [YEN82], [DEG85]. The auto-zero technique is also used to ensure adequate biasing of CMOS inverters or analog-to-digital converters [CAN82]. Dynamic element matching [VdP76] is based on the chopper technique and shifts the error components to higher frequencies. The drawback of this technique is the high residual output ripple, which for most applications must be filtered out by using external components. Furthermore multiple mirrors are difficult to implement. Dynamic analog techniques [VIT85.1] exploit the absence of gate current to temporarily store some analog information on the gate capacitance of the MOS device. A reported application of this analog storage capability is the dynamic comparator [YEE78] which sequentially uses the same transistor as the two devices of a differential pair. With this auto-zero technique the very notion of mismatch disappears. The achievable precision is moved to new limits and depends on the capability of accurately storage the signal, mainly limited by charge injection from the MOS transistors used as switches. Although the idea of current sampling was formulated over ten years ago [OGU78], the first accurate implementation dates from 1988, because the circuit theory and process technology did not allow researchers to obtain the expected results. In the "Electronics Letters" issue of December 1988 [DAU88] published the idea and some simulated results, which induced a series of publications during the year of 1989. Several research laboratories have focused on the subject during the last few years, with the reported results of [GRO89] concerning D/A converters, [NAI89.2] for A/D converters, [HUG89] for current mode circuits and filters, and ourselves [VIT88], [WEG89.1] for dynamic current mirrors. The primary objective being pursued in this dissertation is to investigate the different possibilities, to design and to analyze the performances and limitations of a new type of current mirror, a so-called dynamic mirror or current copier. The goal is to build highly accurate current mirrors in the simplest and most compact way. The best strategy for a given problem can be chosen only if all parameters influencing the circuit performances are known. The outline of this thesis is the following: Chapter 2 presents the principle of memorization of a current copier. The main limitations which influence the achievable accuracy are deduced and cell structures which reduce these effects derived. In Chapter 3 the principle and the different configurations of dynamic current mirrors are extensively described, and their advantages and disadvantages discussed and compared. Possibilities of realizing multiple, multiplying and dividing current mirrors are highlighted. Chapter 4 provides a deeper look at the different parameters, which limit the accuracy of a dynamic current mirror, namely drain voltage variations, leakage currents, noise and charge injection. The sampling of white noise and 1/f noise is analyzed and their contribution calculated. The influence of charge injection is evaluated and the interfering parameters shown. The transient behavior is considered in Chapter 5, where the output spikes and the trade-off between speed and accuracy are highlighted. Chapter 6 emphasizes layout considerations and the practical implementation of a dynamic current mirror. Chapter 7 summarizes the experimental results obtained with such mirrors. Chapter 8 deals with the different possible applications, focusing on a continuous time filter. The principle of D/A & A/D converters and switched current filters are outlined, and the extension to other functional blocs is suggested. Finally Chapter 9 provides summarizing remarks and conclusions.

Fuente: http://infoscience.epfl.ch/record/31368

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

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Circuit Idea / How to Reverse Current Direction

Not a Diode

Article says But it is very primitive and confusing to say "the input transistor Q₁ is a diode". Actually, it is not a diode; it is exactly a transistor operating in the active mode. It would be a diode, if its collector was disconnected. I'd be inclined to notice that the CB junction of the transistor is short-circuited in Q₁. The collector node is attached directly to the base node. Therefore the circuit "sees" only the BE junction, which is, of course, a diode. Sounds like a simple explanation to me. Brews ohare (talk) 00:37, 2 March 2008 (UTC)

Brews ohare, thank you for the response. It is wonderful that there are still (wiki)people showing willingness for discussing circuit phenomena! As you can see, I not only try to "humanize" Wikipedia articles by means of these discussions; I draw also inspiration from these conversations that I need to continue disclosing the mystery of electronic circuits on the pages of Circuit Idea. It will be so marvelous, if more key-wikipedians (e.g., Omegatron, Alfred Centauri, Light current, Heron, Rogerbrent, etc.) join this discussion! Of course, we have to be bold and fair enough for the purpose of such a frank discussion; we have not to hide behind "reputable" sources.

As far as I remember, I have also to reply to your comment in Wikipedia's Current mirror. Below, I will try to expose my viewpoint about this interesting topic.

What Can Be the Current Mirror Building Component?

I'm not familiar enough with microelectronics; I try to look at electronic devices from a macro- instead from a micro-level. From this viewpoint, the specific current mirror implementation is not so interesting for me; I try to "catch" the general idea behind it. Following this approach, let's see what an electronic device we can use as a building component to realize the two parts of the current mirror.

If you share my penetration, the so called transconductance (voltage-to-current conversion) is the main, general and inherent property of the device. It is not so important, if the relation between the output current and the input voltage is linear; but it is important for the device to keep up a steady output current when the load varies. Simply speaking, we need a voltage-controlled current-stable resistor that changes its present resistance contrary to the input voltage: if we raise the input voltage, the resistance decreases and v.v. This property allows us to build the input current-setting (programming, reference) part of the current mirror by applying a negative feedback. For this purpose, we just connect the device's output to its input, in order to reverse it (to transmute it from a voltage-to-current converter into the opposite current-to-voltage converter).

Almost all the electronic devices (tubes, bipolar-, MOS and junction field-effect transistors, the so-called transconductance amplifiers, etc.) are transconductors that control the current flowing through the load (if we want they to produce a voltage, we connect in series a resistor acting as the simplest current-to-voltage converter). But only two of the "conventional" active electronic components (the bipolar transistor and the E-MOS field-effect transistor) possesses the needed input-output relation.


MOSFET as a Current Mirror Building Component

It is exactly what we need since a MOSFET has a true voltage input behaving just as an open circuit. As a result, there are no any problems, if we connect the M1's drain to its gate (Fig. 1). There is no additional current to be extracted from the input source (there is no error because of the input currents); there are no any "diodes" connected in parallel to the device's output... There is only a regulating element (a voltage-controlled resistor) that changes its present resistance (between the drain and the source) so that to keep up an almost constant voltage. We have made it behave in this way - as a voltage-stable resistor, as a forward-biased diode! There are no any other elements in this arrangement...

An E-MOSFET has a threshold in its curve but it affects only the output, not the input (although the threshold voltage is exceeded, the input resistance remains infinite). So, we can think of an E-MOSFET as a kind of "transdiode".




BJT as a Current Mirror Building Component

But BJT is a "nastier" and more confusing device... First, it is a double-faced component; we can think of a bipolar transistor simply as a current-to-current converter (a current amplifier) or, more precisely, as a voltage-to-current converter (according to Ebers-Moll model). Well, we might accept here the second viewpoint... But what do we do with the base-emitter diode? What is its role here? Is it important? Is the transistor T₁ with a shorted CB junction a diode? What is it?

Yes, there is a diode in this arrangement... You are right, the base-emitter junction is really a true diode... But this is only the first diode. Because there is another "diode" - a regulating element (a voltage-controlled resistor) that changes its present resistance (now, between the T1's collector and the emitter) so that to keep up an almost constant voltage. Again, we have made it behave in this way - as a voltage-stable resistor, as another forward-biased diode!

So, in this arrangement, we have two diodes - the first (base-emitter junction) is genuine, the second (collector-emitter part) is artificial. And these two diodes are connected in parallel to each other. Only, the BE diode is low-power (driving) while the CE "diode" is powerful, "executive". The BE diode consumes only 1/(1 + β) portion of the whole current. The CE part serves as a shunting regulating element that diverts the great amount (β/(1 + β)) of the current. So, what the circuit "sees" is not only a diode, it "sees" a shunted, powerful diode... We can even say that since the CE shunting "diode" predominates over the BE true diode (it is β times more powerful than the BE diode) the circuit "observes" only the CE "diode"!

Well, Brews ohare, that's enough; I suggest correcting the original article text from
...But it is very primitive and confusing to say "the input transistor Q₁ is a diode". Actually, it is not a diode; it is exactly a transistor operating in the active mode. It would be a diode, if its collector was disconnected...

to
...But it is not sufficient to say "the input transistor Q₁ is a diode" since it is not only a diode; it is a "buffered diode". It would be a bare diode, if its collector was disconnected and only the base-emitter junction was connected...
or to something similar.



True Diode as a Current Mirror Input Part

Brews ohare, I would like to ask you to take up a position on another "scenario" for building BJT current mirror. In this story, we may first use a true diode as a current-setting input part and, as usual, a BJT as an output part. Then, we will note some disadvantages of this arrangement that will make us use another BJT as an input part.

This idea dawned on me when I, looking for current mirror materials on the web, found the Tony Kuphald's Current mirrors page. He has used such a succession; only he has not given reasons for such a circuit evolution. I regard with respect his web materials; IMO they are the best written popular circuit stories. I know him and I have the intention of inviting him to join Circuit idea stories.

Only, I think there is something confusing and wrong in his material. Can you read thoroughly this material and then give your opinion on it? Do you see something wrong in this arrangement - to connect a bare diode in parallel to the T2's input? Can such an arrangement exist at all? Can it serve as a current mirror?

If we answer these questions, we can show in another credible way the reason of using an active diode. Then, we might invite Tony Kuphaldt to join this discussion, to edit this page or to write new one.

I have an opinion but I prefer first to see what you think about it.
Regards, Circuit-fantasist (talk) 18:35, 2 March 2008 (UTC)

Vout of Input Part is Const

I'm confusing about this point: If this Vout of input part is a const and won't be changed with the input currrent, then the output current won't be changed also as this voltage is Vbe of output part and therefore will decide the output current. So the result is if we change the input current configuration resistor, the output current won't follow this change;
Would you please give me a more detail explanation? Thanks!

Hi, your question is interesting and important. Vout (or V_F of an ordinary diode) is relatively constant. In some cases, e.g. in a case of a voltage stabilizer or if the second device has a large operating range, we assume it is constant because its variations are insignificant in comparison with the voltage range. In the case of a current mirror, we assume it depends logarithmically on the input current in comparison with the transistor's input voltage range. Circuit-fantasist (talk) 16:03, 15 February 2009 (UTC)

The FB-Based Reverse is Incomplete

It is amazing how one can decompose an initially miraculous circuit into principal blocks. Yet, thinking in terms of feedback causes trouble when trying to understand the metamorphosis turning the transistor into its inverse. This book helped me to understand that the feedback not only a way to keep the output at specific level but also a lookup of the input that produces given output. Normally, "internal input" i is adjusted by i = i+Δi, where Δi = ref - o is the difference between primary input ref and "internal output" o. In your example, the current (and collector voltage) is the reference input and base voltage is the internal input. But I see no comparator. Instead, I see that the internal output controls our golden reference, the input current! You see, instead of matching with the golden signal, feedback controls it. Next, the two drivers, the primary input and our transistor converter, must fight -- who is stronger. Analysis reveals that the mirror reference input always wins because, when the transistor is too closed, the current can go through the base (and open it). It will also close the transistor if the collector voltage is too low for the given current. The solution comes from understanding the transistor operation while the feedback view seems only to cause trouble. The book is otherwise great, yet, I feel like it lacks a last remark to explain the trick of actual FB implementation. Or, is it supposed that the audience is of competence to guess itself? --Javalenok (talk) 14:39, 17 October 2009 (UTC)

Javalenok, your thoughts are very interesting for me. I noted your insertion this evening; so, please let mi scrutinize them. If you want, you may join as well another interesting talk - Wikipedia discussion about Emitter-coupled logic where I appear as Circuit-dreamer and sometimes with my old WP user name Circuit-fantasist. Regards, Circuit-fantasist (talk) 20:10, 21 October 2009 (UTC)

Javalenok, I have found a very interesting source (page 12 and 16) that seconds this current mirror presentation. Circuit-fantasist (talk) 22:40, 24 October 2009 (UTC)

Fuente: http://en.wikibooks.org/wiki/Talk:Circuit_Idea/How_to_Reverse_Current_Direction

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

 

A High Speed Current Mirror Memory Cell Architecture

Background

Owing to its higher electron mobility, gallium arsenide (GaAs) is thought to have potential for outperforming silicon devices in digital integrated circuits. To achieve high performance, GaAs must be able to incorporate adequate amounts of high-speed memory on chip. High speed GaAs microprocessors developed to date have integrated only small amounts of memory on chip. Future designs will require large sub-2 ns on-chip caches.

Technology Description

University of Michigan researchers have developed a new memory cell architecture for GaAs MESFETs. This access current increase improves access time by about 25-50% in memory arrays of 4kb to 16kb. This new cell is immune to the possibility of destructive read associated with conventional cells. The biasing of this cell minimizes the impact of leakage currents on the number of bits that can be safely connected to a column. Measurements indicate this SRAM can work with up to 512 bits per column for 75Y'C operation. In addition to having faster read times, this cell can be written to much faster than conventional memory cells. A memory has been simulated and designed around this new cell, and preliminary tests have been conducted. The University seeks to license this technology for the development and commercialization of products.

Applications

• On-chip cache memories, test equipment,graphics and signal processors

Advantages

• Faster read and write speeds
• Removes the write after read hazardof the conventional cell architecture
• Eliminates the need to limit bit-lineswing during read

Fuente: http://www.ibridgenetwork.org/umich/a-high-speed-current-mirror-memory-cell-architecture?search_term=Current+Mirror+

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

Circuit Idea/How the Wilson Current Mirror Keeps the Current

The Two Aspects of the Wilson Current Mirror

We may consider the behavior of Wilson current mirror from two aspects. From one hand, when we vary the input quantities (the input voltage, the resistance R or actually, the input current), a current mirror behaves as a current follower. In this case we have considered in the previous story, it is important the output current to follow exactly the input current. From the other hand, if we vary the output quantities (the supply voltage, the load resistance or voltage), the current mirror behaves as a constant current source. In this case, it is important the circuit to keep up a steady current. Well, let's discuss the second aspect here.

The Second Problem of the Simple Current Mirror

The output part of the simple BJT current mirror exploits the basic property of the bipolar transistor to behave as a current-stable resistor if we keep up steady its base voltage or current. Actually, in combination with the power supply, the transistor constitutes a current source or sink. How does the transistor do this magic? If, for example, the load resistance R_L varies, the transistor changes its present resistance R_T between the collector and the emitter so that to keep up a constant total resistance R_tot = R_L + R_T = const (Fig. 1).

Only, due to Earley effect, the output part does not behave as a perfect current source; this is the second imperfection of the simple current mirror that we have to improve now.



Making the Transistor Keep a Constant Current by Negative Feedback

We can make the output transistor keep up a constant current by applying various clever "tricks" but negative feedback is the more reliable of them. Then, how do we introduce such a "current-keeping" negative feedback in the simple transistor stage? Let's begin thinking...

We have a transistor that controls the current through the load by changing its present resistance between the collector and the emitter... What does it mean to introduce a "current-keeping" negative feedback? What does have to happen if the current tries to change because of R_L or V_CC variations? Obviously, if the load current tries to increase, the transistor has to close more (to increase its present collector-emitter resistance R_T) so that to restore the previous magnitude of the current. And v.v., if the load current tries to decrease, the transistor has to open more (to decrease its present collector-emitter resistance) - Fig. 2. For simplicity, let's consider only the first case (the load current increases) from now on.

In order to make the transistor do this magic, we have to close the negative feedback loop from the output circuit where the load current flows to the transistor input. What is the transistor input? The base-emitter junction (gate-drain part) serves as a differential voltage input; but, regarding to the ground, the transistor has two single-ended voltage inputs - the emitter and the base. So, we might introduce two kinds of negative feedbacks by applying a voltage that is proportional to the load current to the emitter and to the base.



...Driving the Transistor from the Emitter...

First, we may fix the base voltage and to drive the transistor from the emitter. For this purpose, we connect a reference voltage source V_REF to the base and a current-to-voltage converter in the emitter producing a voltage proportional to the load current I_L (Fig. 3).



A bare resistor R_E can serve as a simple current-to-voltage converter (Fig. 4). Since the output quantity (the voltage drop V_Re) is applied in series to the input quantity (the voltage V_REF) this popular technique for keeping up a constant current is named series negative feedback or "emitter degeneration".
Only, the voltage drop V_Re limits the maximum voltage drop across the load (the so called voltage appliance). What do we do then?



...Driving the Transistor from the Base

General Idea

But don't you think that, with the same success, we may fix the emitter voltage and drive the transistor from the base? Let's try it! Maybe, it will lead us to the desired Wilson current mirror... In order to impelement this idea, we need again an element that produces a voltage proportional to the load current, i.e. current-to-voltage converter. But now the current flows in one place (the emitter) while the voltage has to be applied to other place (the base)! So, we need not a bare "resistance" I-to-V converter; we need a kind of "transresistance" I-to-V converter. How do we make it? How have Wilson solved this problem?

Eureka! We may "copy" the load current I_L to the desired place where to pass it through a resistor, in order to create a voltage drop proportional to I_L (Fig. 5).



Then let's do it! For this purpose, we connect an I-to-V converter in the emitter that drives a reverse V-to-I converter. As you can see, the direct and the reversed converter constitute actually the well-known simple current mirror. It produces a "copy" of I_L that flows through a conventional resistive I-to-V converter connected to the base (Fig. 6). In this way, by means of a current mirror and a current-to-voltage converter we apply again a current-keeping negative feedback.

The Wilson current mirror consists of a simple current mirror and a current-to-voltage converter connected in the feedback loop.



Implementation

As above, a bare resistor R can act as a simplest current-to-voltage converter (Fig. 7). Actually, it serves as the emitter "degeneration" resistor R_E from Fig. 4. Only, here not the "original" load current I_L flows through the resistor R but a "copy" I_R = I_L of this current. The voltage drop V_R across the resistor R or, more strictly speaking, its supplemental to V_CC voltage is the input voltage for the transistor T₃



Operation

In order to ascertain if this odd circuit behaves as a constant current source, we have somehow to "provoke" it and to see its reaction to our "intervention". Well, how may we "provoke" it? Let's for instance, investigate how the circuit will react if we change the load resistance R_L. In the beginning, suppose equal currents I_L = I_R flow through the two circuit legs what is normal circuit condition.

If we increase R_L, the load current I_L tries to decrease. This current is the input quantity of the simple current mirror T₁, T₂; so, its output quantity I_R decreases also. As a result, the voltage drop V_R decreases and its supplement V_CE2 increases. The input voltage V_BE3 of the transistor T₃ increases; it begins opening more until the load current restores its previous magnitude.

What does the transistor T₃ actually keep? It keeps up a constant base-emitter voltage V_BE. If we look at this circuit as a negative feedback stabilizer, V_BE is its input reference quantity and the transistor T₃ keeps constant this quantity. Doing that, it keeps actually a constant voltage drop V_R across the steady resistor R; so, the current I_R and I_L are constant too

How Many Feedbacks are There?

The most resources about Wilson current mirror do not notice any feedback in the circuit. Some of them have noticed two positive feedbacks, other - two negative ones. But actually there are as many as three negative feedbacks in this sophisticated circuit! Let's try to see them.

1. A Local Feedback - the wire between the T₁'s collector and its base. It reverses T₁ making it act as a logarithmic current-to-voltage converter instead as an antilogarithmic voltage-to-current converter what is inherent for BJT. This is a parallel constant-voltage keeping negative feedback.

2. A More Global Feedback - the whole active diode (T₁ + the wire between its collector and the base) that is connected in the T₃'s emitter. This is a series constant-current keeping feedback (an emitter degeneration). Only, it is negligible for keeping the constant current because the active diode behaves as a voltage-stable element (significant current variations cause insignificant voltage changes).

3. A Global Negative Feedback - it is constituted by the simple current mirror (T₁ and T₂) and the resistor R acting as a simple current-to-voltage converter. It is the important negative feedback that keeps the current constant; we have already discussed it above.

Fuente: http://es.wikipedia.org/wiki/Espejo_de_corriente

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

 

Circuit Idea / How the Wilson Current Mirror Equalizes the Currents

How do we reveal the secrets of Wilson current mirror?

The Great Challenge

It was a challenge to reveal the idea behind the popular BJT current mirror; but it is a great challenge to disclosure the mystery of the legendary Wilson current mirror (Fig.)! Maybe, there is no so simple (containing only three transistors) and, at the same time, so incomprehensible and misunderstood circuit as Wilson current mirror. There are many resources that have tried to explain this sophisticated, ingenious and elegant circuit solution by using formal methods. But they do not give us what we need, first and foremost, as human beings - the basic idea(s) behind this odd, strange and exotic circuit. It is a great paradox to calculate circuit without knowing the basic idea behind it!?! So, before showing in detail how to calculate the electronic circuit we have first to show what the very basic idea is behind the circuit.

Questions to be Answered

Looking at the circuit diagram (Fig.), we need to answer dozens of questions that are never answered.



What does the transistor Q₃ do in this circuit? What is its function there? Why this current mirror contains another simpler current mirror Q₁ and Q₂ (why an additional simple current mirror is nested in the main current mirror)?!?! What is its function? But why this simple current mirror is reversed (why the transistors Q₁ and Q₂ are swapped)? Why the Q₁'s collector current serves as an input quantity and the Q₂'s collector current as an output one (we thought the Q₂'s collector current was the input quantity and the Q₁'s collector current was the output quantity)? Are there negative feedbacks in Wilson current mirror? If there are, what are they? How many negative feedbacks there are? What are their functions (why they are included)? What and how do they control - voltage or current? What are the advantages of Wilson current mirror versus current mirror with emitter degeneration? How does Wilson current mirror keep up an almost constant current (why it has an almost infinite output resistance)? Why the input and output currents are almost equal (what is the trick)? Has a MOSFET Wilson current mirror some advantage versus the simple MOSFET current mirror?

Heuristic Approach

Since there are no satisfactory answers to the questions above, let's try to disclose the mystery of the famous circuit by ourselves; let's answer these questions relying mainly on our human intuition, imagination and common sense. Please, just forget all kinds "cut-and-dried" citations and begin thinking by yourself to make an exciting discussion here!

The best way of understanding and presenting electronic circuits is by reinventing them, by showing the circuit evolution. So, let's imagine how Wilson has invented his current mirror by reinventing and building it, in order to grasp the basic ideas behind the circuit and then to present them in an attractive manner to readers. Of course, it would be wonderful if the very Wilson, if he is still alive, would expose how he has invented the famous circuit. Only, it is a well-known truth that, as a rule, due to variety of reasons, inventors do not show willigness for disclosing the process of invention.

The Imperfections of the Simple Current Mirror

BJT. The simple BJT current mirror (Fig) has two main imperfections: first, the output current differs from the input one because of the two base currents that the transistors Q₁ and Q₂ "suck" from the input current; second, the output current varies when the output (load) voltage changes because of the Early effect. These are completely different problems; there is no any connection between them. Wilson was a lucky man; in the early 60's, he "killed two birds with one stone" - adding only one transistor to the humble current mirror, he managed to remove the both imperfections of the simple BJT current mirror!

MOSFET. The simple MOSFET current mirror has only the second imperfection because there are no gate currents. And even in this case the Wilson idea is beneficial; it enables the MOSFET Wilson current source to keep up a constant output current (see another story about the famous circuit).



How the Wilson Current Mirror Equalizes the Input and Output Currents

In order to justify the name current mirror, in this circuit the output current has to follow exactly the input current. So, regarding to the current magnitudes, a current mirror is actually a current follower. The Wilson current mirror meets closely this requirement. Here we will show how the Wilson current mirror eliminates the difference between the input and output currents.

Posing the First Problem of the BJT Simple Current Mirror

In our routine, we frequently need to equalize two quantities having different magnitudes. Let's, for examaple, scrutinize a popular mechanical analogy: imagine two heaps of weights are placed on the two sides of ordinary mechanical scales - for concreteness, 4 grams on the left pan and 6 grams on the right pan Fig.



Since there is a difference of two grams between them (4 < 6), the scales are unbalanced (inclined to the left). This mechanical analogy corresponds to the circuit of the BJT simple current mirror (Fig. A) consisting of transistors with ß = 4 (Fig. B) if we assume that one base current corresponds to one gram. The problem is how to equalize the two weights, respectively the two currents.

More generally and more precisely speaking, we want to create a copy of an original quantity but two entities are lost from the original before we make the copy. The problem is how to compensate the loss.



For this purpose, let's try to present the ideas behind this legendary circuit and its operation in a more attractive way. We know, in order to grasp the ideas behind circuits, we may visualize electrical quantities voltage and current by voltage bars and current loops. In the case of current mirrors, it is extremely interesting to show where currents flow and to visualize their magnitudes. So, we may draw two kinds of figures in this section. The first, placed on the left side of the page, will represent the respective circuit diagram with superimposed voltage and current "maps". The second, placed on the right side of the page, will visualize the magnitudes of the currents by fat lines whose thickness is proportional to the magnitude of the corresponding current.
Let's begin with presenting the simple BJT current mirror in such an attractive way (Fig). You can see on the left picture (Fig. A) where currents flow and particularly how the transistors "suck" two base currents. The currents are represented by closed loops (every current finishes where it has started). The lines on the right picture (Fig. B) are actully sections of the closed current loops from Fig. A. The transistors are shown with extremely low ß (on this figure ß = 4) for convenience, in order to present the base currents by thick enough lines.



Equalizing the Currents by "Pushing" 2I_B to I_IN...

How do we solve the poblem of the two "sucking" I_B in the simple BJT current mirror? Let's begin thinking relying on our human common sense...
We know from our daily routine the great idea of compensation: if there are some losses of "something", we might compensate them by adding the same quantity of this "thing". In our mechanical analogy, we may just take two grams from the weight box and add them to the left pan, in order to balance the scales (4 + 2 = 6) - Fig.

Generally speaking, here we compensate the loss by adding two items to the original (we correct completely the original).



Well, let's put in practice this simple idea! "Electrically" speaking, since the transistors "suck" two I_B, the first idea that might dawn on us is, of course, to add the same two I_B (Fig. B). Then I_OUT = I_IN - 2I_B + 2I_B = I_IN. For this purpose, we have to connect an injecting current source 2I_B to the T₁'s collector - Fig. A (more precisely speaking, this is rather a current-stable resistor than a source).

Only, this has to be not an ordinary constant current source but a "following" current source that copies the current 2I_B. What an idiocy! It turns out that we need another current mirror?!?






...By "Sucking" 2I_B from I_OUT...

With the same success, in our mechanical analogy, we may subtract two grams from the right pan and place them to the weight box, in order to balance the scales again (4 = 6 - 2) - Fig. So, we may reason in similar way about the circuit...

Generally speaking, now we compensate the loss by adding two items to the copy (we correct completely the copy).




Let's now apply this "mechanical" idea to our electrical circuit... Above, we have added the two compensating base currents by injecting them into the input current (the "original" quantity). But with the same success we might add them to the output current (the "copy") by "sucking" 2I_B from it (Fig. B). Now I_IN = I_OUT - 2I_B + 2I_B = I_OUT. In this case, we have to connect a sinking current source 2I_B to the T₂'s collector (Fig. A).

As before, this has to be not an ordinary constant current source but a "following" current source that copies the current 2I_B.






...By Both "Sucking" I_B from I_OUT and "Pushing" I_B to I_IN...

We have almost reached the great Wilson's current equalizing idea... Well, let's continue thinking. It is inconvenient to create 2I_B; it is easier to produce (source or sink) only I_B. What do we do then?

Let's go back again to our favorite mechanical analogy; it may help us... And really, instead to add as many as two grams to the left pan or to subtract two grams from the right pan, we may balance the scales by adding only one gram from the weight box to the left pan and subtracting one gram from the right pan placing it to the weight box (4 + 1 = 6 - 1) - Fig.

Generally speaking, now we compensate the loss by adding one item to the original and another item to the copy (we compensate partially both the original and the copy).



Let's now put in practice this interesting idea... For this purpose, we might connect an injecting current source I_B between the positive rail and the T₁'s collector and a sinking current source I_B between the T₂'s collector and the ground - Fig. 10a. In this way, we add one base current to the input current and another base current to the output current (Fig. 10b). As a result, the two currents become equal:


I_IN + I_B = I_OUT + I_B, so, I_IN = I_OUT






...By "Moving" I_B from I_OUT to I_IN...

We have really reached the great Wilson's current equalizing idea... We may develop further the powerful idea from the mechanical analogy above just moving one gram from the right pan to the left pan! Well, let's say it again: instead to add one gram to the left pan and to subtract one gram from the right pan, we may balance the scales just by moving one gram from the left to the right pan (4 + 1 = 6 - 1) - Fig. But this is not only Wilson's idea! Every vendor from the past knew and used this clever trick!

Generally speaking, now we compensate the loss by moving one item from the original (before the loss) to the copy. Thus we compensate partially both the original and the copy; we redistribute the compensating items.




Let's now put in practice this clever trick... As above, we add one base current to the input current and another base current to the output current (Fig. B). As a result, the two currents become again equal:
I_IN + I_B = I_OUT + I_B, so, I_IN = I_OUT   Here is the great Wilson's idea! In order to equalize the two currents, he has just "moved" one base current from the one to the other leg!
Don't you think this connection resembles a bridge circuit (the current source serves as a "bridge" between the two circuit legs)?





Realizing the Wilson's Current Equalizing Idea

Once we revealed the brilliant Wilson's idea we have only to implement it. Let's continue thinking...

What is this mysterious element that can consume I_B from one part of the circuit and can add it to the other part? Of course, there is nothing more natural for a bipolar transistor to do that "donkey work"! It "sucks" I_B from the point where its base is connected and adds it to the emitter current. Then let's connect a transistor T₃ in the left leg of our circuit (Fig. A)!

Wonderful, now it sinks the current I_B from I_OUT and injects the same current I_B into I_IN (Fig. B)! So, we have managed to reveal the role of the mysterious transistor T₃!

In the circuit of Wilson current mirror, the transistor T₃ "moves" one base current from the right to the left leg of the circuit.





Reversing the Circuit to Obtain the True Wilson Current Mirror

Only, there is something wrong in this connection because we can't change the input current in the left leg (by varying the resistor or the voltage). If we try to do that, the transistor T₃ will resist to our intervention thus keeping up a constant current (see another story about the legendary circuit). Let's continue thinking.

Well, the transistor T₃ has to adjust its base current so that its collector current to remain unchanged as we want. The only "thing" that can do this magic is the ubiquitous negative feedback that keeps up an almost constant voltage (this should be the same kind of negative feedback as this applied to the transistor T₁).

But there isn't a feedback in this input part of the circuit; there is no connection between the T₃'s collector and base. Instead, there is a negative feedback implemented by T₃ and T₁ that are connected between the collector and the emitter of T₂ in the output part of the circuit (see again Fig. A)! What do we do then?

Eureka! We may swap the two circuit legs: the output part can serve as an input one and the input part - as an output one. Thus we obtain finally a true Wilson current mirror (Fig. A)! Let's draw the final conclusion:

In the circuit of Wilson current mirror, the transistor T₃ "moves" one base current from the left to the right leg of the circuit.

Fuente: http://en.wikibooks.org/wiki/Circuit_Idea/How_the_Wilson_current_mirror_equalizes_the_currents

Nombre: Rodriguez B. Joiver I.

Asignatura: EES

 

A High Precision CMOS Current Mirror / Divider

This mirror circuit, for use in analog and mixed-signal integrated circuits, provides very high precision and a wide range of current divisions.

Suggested Uses

This mirror circuit is for applications such as high accuracy A/D and D/A converters, reference cells, and high current comparators.

Advantages

Current mirror circuits, common to analog and mixed-signal circuits offer wide bandwidth and have been developed in bipolar, MOS and BiCMOS technologies. Improvements in their designs have been numerous, each enhancement aimed at specific problems and its application. Generally, in CMOS technology, the dependency of the output current on the voltage strongly affects both current mirror configurations. Also the up mirror depends upon the P transistors and the down mirror depends upon the N transistors, each with different characteristics which give rise to matching problems of the two mirrors. In this invention, the performances of the two current mirror topologies are matched. Thus, compared to other topologies, the circuit topology of this invention offers several notable advantages which include ease of design, close to ideal up and down mirroring, insensitivity to power supply variations of the up and down mirrored currents and good operational insensitivity to process parameter variations, thereby requiring no trimming or self-calibration.

Innovation Details

Detailed Description

This invention relates to current mirror circuits useful especially in analog and mixed-signal integrated circuits and particularly to a current mirror circuit that provides very high precision and a wide range of current divisions, for applications such as high accuracy A/D and D/A converters, reference cells, and high current comparators.
Current mirror circuits, common to analog and mixed-signal circuits offer wide bandwidth and have been developed in bipolar, MOS and BiCMOS technologies. Improvements in their designs have been numerous, each enhancement aimed at specific problems and its application. Generally, in CMOS technology, the dependency of the output current on the voltage strongly affects both current mirror configurations. Also the up mirror depends upon the P transistors and the down mirror depends upon the N transistors, each with different characteristics which give rise to matching problems of the two mirrors. In this invention, the performances of the two current mirror topologies are matched. Thus, compared to other topologies, the circuit topology of this invention offers several notable advantages which include ease of design, close to ideal up and down mirroring, insensitivity to power supply variations of the up and down mirrored currents and good operational insensitivity to process parameter variations, thereby requiring no trimming or self-calibration.

Cascode

The Cascode is a two-stage amplifier composed of a transconductance amplifier followed by a current buffer. Compared to a single amplifier stage, this combination may have one or more of the following advantages: higher input-output isolation, higher input impedance, higher output impedance, higher gain or higher bandwidth. In modern circuits, the cascode is often constructed from two transistors (BJTs or FETs), with one operating as a common emitter or common source and the other as a common base or common gate. The cascode improves input-output isolation (or reverse transmission) as there is no direct coupling from the output to input. This eliminates the Miller effect and thus contributes to a much higher bandwidth.

History

The cascode (sometimes verbified to cascoding) is a universal technique for improving analog circuit performance, applicable to both vacuum tubes and transistors. The word "cascode" is a contraction of the phrase "cascade to cathode". It was first used in an article by F.V. Hunt and R.W. Hickman in 1939, in a discussion for application in low-voltage stabilizers. They proposed a cascode of two triodes (first one with common cathode, the second one with common grid) as a replacement of a pentode.

Operation

Figure shows an example of cascode amplifier with a common source amplifier as input stage driven by signal source V_in. This input stage drives a common gate amplifier as output stage, with output signal V_out.



The major advantage of this circuit arrangement stems from the placement of the upper Field Effect Transistor (FET) as the load of the input (lower) FET's output terminal (drain). Because at operating frequencies the upper FET's gate is effectively grounded, the upper FET's source voltage (and therefore the input transistor's drain) is held at nearly constant voltage during operation. In other words, the upper FET exhibits a low input resistance to the lower FET, making the voltage gain of the lower FET very small, which dramatically reduces the Miller feedback capacitance from the lower FET's drain to gate. This loss of voltage gain is recovered by the upper FET. Thus, the upper transistor permits the lower FET to operate with minimum negative (Miller) feedback, improving its bandwidth.

The upper FET gate is electrically grounded, so charge and discharge of stray capacitance C_dg between drain and gate is simply through R_D and the output load (say R_out), and the frequency response is affected only for frequencies above the associated RC time constant: τ = C_dg R_D//R_out, namely f = 1/(2πτ), a rather high frequency because C_dg is small. That is, the upper FET gate does not suffer from Miller amplification of C_dg.

If the upper FET stage were operated alone using its source as input node (i.e. common-gate (CG) configuration), it would have good voltage gain and wide bandwidth. However, its low input impedance would limit its usefulness to very low impedance voltage drivers. Adding the lower FET results in a high input impedance, allowing the cascode stage to be driven by a high impedance source.

If one were to replace the upper FET with a typical inductive/resistive load, and take the output from the input transistor's drain (i.e. a common-emitter (CE) configuration), the CE configuration would offer the same input impedance as the cascode, but the cascode configuration would offer a potentially greater gain and much greater bandwidth.

Stability

The cascode arrangement is also very stable. Its output is effectively isolated from the input both electrically and physically. The lower transistor has nearly constant voltage at both drain and source and thus there is essentially "nothing" to feed back into its gate. The upper transistor has nearly constant voltage at its gate and source. Thus, the only nodes with significant voltage on them are the input and output, and these are separated by the central connection of nearly constant voltage and by the physical distance of two transistors. Thus in practice there is little feedback from the output to the input. Metal shielding is both effective and easy to provide between the two transistors for even greater isolation when required. This would be difficult in one-transistor amplifier circuits, which at high frequencies would require neutralization.

Biasing

As shown, the cascode circuit using two "stacked" FET's imposes some restrictions on the two FET's―namely, the upper FET must be biased so its source voltage is high enough (the lower FET drain voltage may swing too low, causing it to leave saturation). Insurance of this condition for FET's requires careful selection for the pair, or special biasing of the upper FET gate, increasing cost. The cascode circuit can also be built using bipolar transistors, or MOSFETs, or even one FET (or MOSFET) and one BJT. In the latter case, the BJT must be the upper transistor; otherwise, the (lower) BJT will always saturate (unless extraordinary steps are taken to bias it).

Advantages

The cascode arrangement offers high gain, high slew rate, high stability, and high input impedance. The parts count is very low for a two-transistor circuit.

Disadvantages

The cascode circuit requires two transistors and requires a relatively high supply voltage. For the two-FET cascode, both transistors must be biased with ample V_DS in operation, imposing a lower limit on the supply voltage.

Dual-Gate Version

A dual-gate MOSFET often functions as a "one-transistor" cascode. Common in the front ends of sensitive VHF receivers, a dual-gate MOSFET is operated as a common-source amplifier with the primary gate (usually designated "gate 1" by MOSFET manufacturers) connected to the input and the 2nd gate grounded (bypassed). Internally, there is one channel covered by the two adjacent gates; therefore, the resulting circuit is electrically a cascode composed of two FETs, the common lower-drain-to-upper-source connection merely being that portion of the single channel that lies physically adjacent to the border between the two gates.

Other Applications

With the rise of integrated circuits, transistors have become cheap in terms of silicon die area. In MOSFET technology especially, cascoding can be used in current mirrors to increase the output impedance of the output current source. A modified version of the cascode can also be used as a modulator, particularly for amplitude modulation. The upper device supplies the audio signal, and the lower is the RF amplifier device.

Two-Port Parameters

The cascode configuration can be represented as a simple voltage amplifier (or more accurately as a g-parameter two-port network) by using its input impedance, output impedance, and voltage gain. These parameters are related to the corresponding g-parameters below. Other useful properties not considered here are circuit bandwidth and dynamic range.

BJT Cascode: Low-Frequency Small-Signal Parameters

The idealized small-signal equivalent circuit can be constructed for the circuit in figure by replacing the current sources with open-circuits and the capacitors with short circuits, assuming they are large enough to act as short-circuits at the frequencies of interest. The BJTs can be represented in the small-signal circuit by the hybrid-pi model.



MOSFET Cascode: Low-Frequency Small-Signal Parameters

Similarly the small-signal parameters can be derived for the MOSFET version, also replacing the MOSFET by its hybrid-pi model equivalent. This derivation can be simplified by noting that the MOSFET gate current is zero, so the small-signal model for the BJT becomes that of the MOSFET in the limit of zero base current:

,

Where V_T is the thermal voltage.



The combination of factors g_mr_O occurs often in the above formulas, inviting further examination. For the bipolar transistor this product is (see hybrid-pi model):

.

In a typical discrete bipolar device the Early voltage V_A ≈ 100 V and the thermal voltage near room temperature is V_T ≈ 25 mV, making g_mr_O ≈ 4000, a rather large number. From the article on hybrid-pi model, we find for the MOSFET in the active mode:

At the 65 nanometer technology node, I_D ≈ 1.2 mA/μ of width, supply voltage is V_DD = 1.1 V; V_th ≈ 165 mV, and V_ov = V_GS-V_th ≈ 5%V_DD ≈ 55 mV. Taking a typical length as twice the minimum, L = 2 L_min = 0.130 μm and a typical value of λ ≈ 1/(4 V/μm L), we find 1/λ ≈ 2 V, and g_mr_O ≈ 110, still a large value. The point is that because g_mr_O is large almost regardless of the technology, the tabulated gain and the output resistance for both the MOSFET and the bipolar cascode are very large. That fact has implications in the discussion that follows.

Low Frequency Design

The g-parameters found in the above formulas can be used to construct a small-signal voltage amplifier with the same gain, input and output resistance as the original cascode (an equivalent circuit). This circuit applies only at frequencies low enough that the transistor parasitic capacitances do not matter. The figure shows the original cascode (top panel) and the equivalent voltage amplifier or g-equivalent two-port (bottom panel). The equivalent circuit allows easier calculations of the behavior of the circuit for different drivers and loads. In the figure a Thévenin equivalent voltage source with Thévenin resistance R_S drives the amplifier, and at the output a simple load resistor R_L is attached. Using the equivalent circuit, the input voltage to the amplifier is (see article on voltage division):

,

Which shows the importance of using a driver with resistance R_S << R_in to avoid attenuation of the signal entering the amplifier. From the above amplifier characteristics, we see that R_in is infinite for the MOSFET cascode, so no attenuation of input signal occurs in that case. The BJT cascode is more restrictive because R_in = r_π2.

In a similar fashion, the output signal from the equivalent circuit is

,

In low frequency circuits, a high voltage gain typically is desired, hence the importance of using a load with resistance R_L >> R_out to avoid attenuation of the signal reaching the load. The formulas for R_out can be used either to design an amplifier with a sufficiently small output resistance compared to the load or, if that cannot be done, to decide upon a modified circuit, for example, to add a voltage follower that matches the load better.

The earlier estimate showed that the cascode output resistance is very large. The implication is that many load resistances will not satisfy the condition R_L >> R_out. (An important exception is driving a MOSFET as load, which has infinite low frequency input impedance.) However, the failure to satisfy the condition R_L >> R_out is not catastrophic because the cascode gain also is very large. If the designer is willing, the large gain can be sacrificed to allow a low load resistance; for R_L << R_out the gain simplifies as follows:

.

This gain is the same as that for the input transistor acting alone. Thus, even sacrificing gain the cascode produces the same gain as the single-transistor transconductance amplifier, but with wider bandwidth.

Because the amplifiers are wide bandwidth, the same approach can determine the bandwidth of the circuit when a load capacitor is attached (with or without a load resistor). The assumption needed is that the load capacitance is large enough that it controls the frequency dependence, and bandwidth is not controlled by the neglected parasitic capacitances of the transistors themselves.



High Frequency Design

At high frequencies, the parasitic capacitances of the transistors (gate-to-drain, gate-to-source, drain-to body, and bipolar equivalents) must be included in the hybrid pi models to obtain an accurate frequency response. The design goals also differ from the emphasis on overall high gain as described above for low-frequency design. In high frequency circuits, impedance matching at the input and output of the amplifier is typically desired in order to eliminate signal reflections and maximize power gain. In the cascode, the isolation between the input and output ports still is characterized by a small reverse transmission term g₁₂, making it easier to design matching networks because the amplifier is approximately unilateral.

Fuente: http://en.wikipedia.org/wiki/Cascode

Nombre: Rodriguez B. Joiver I.

Asignatura: EES